Quadrature demodulation with phase shift

ABSTRACT

A quadrature demodulator preweights an input signal prior to mixing with in-phase and quadrature clock signals. In an implementation with discrete phase rotation, a series of weighting circuits may be arranged before or after a select circuit to select the amount of phase rotation. Various implementations may include ratioed current mirrors to perform the weighting function, a stacked arrangement of mixers, an H-bridge input stage, integrated mixers and select circuits, and/or selectable gain stages such as gm cells to perform the weighting function.

RELATED APPLICATION

This application claims priority and is a continuation application ofU.S. patent application Ser. No. 11/061,366, titled QuadratureDemodulation with Phase Shift, filed Feb. 17, 2005, which isincorporated by reference.

BACKGROUND

The Doppler effect is widely used in monitoring systems to measure thespeed of moving objects. Some examples include measurement of blood flowwith medical ultrasound equipment and determination of aircraft movementwith radar. The Doppler effect causes the frequency of a wave reflectedfrom a moving object to shift relative to the frequency of the wavedirected at the object. The amount of frequency shift is determined bythe speed of the object.

FIG. 1 illustrates a basic system for measuring Doppler frequency shift.A master oscillator 10 generates a reference frequency signal that isamplified by transmit (Tx) amplifier 12 to generate an electrical signalthat drives a transmit element 14. The transmit element emits waves thatare reflected by an object of interest 16 and return to a receive (Rx)element 18 where they are converted to electrical signals. A receiveamplifier 20 boosts the power of the Rx signal which is then demodulatedby a mixer 22 that mixes the Rx signal with a reference signal derivedfrom the master oscillator. A filter stage 24 having high-pass andlow-pass filters (HPF and LPF) removes unwanted components from thedemodulated signal before further processing, typically by a digitalsignal processing system.

The transmit and receive elements are often mounted together in a singletransducer. In a radar system, the transmit element may be embodied asan antenna which converts the electrical signals into electromagneticwaves. The receive element then converts the reflected electromagneticwaves back into electrical signals. In an ultrasound system, thetransmit and receive elements may be realized as crystals which convertelectrical signals into sound waves and vice versa.

The system of FIG. 1 is known as a coherent demodulation system becausethe reference signal for the demodulator is derived from the same masteroscillator used to generate the transmit signal. It is a non-directionalsystem because it cannot distinguish between motion towards thetransducer and motion away from the transducer. It can only determinethe magnitude (speed) of the motion. This is because motion of theobject towards and away from the transducer create upper and lowerDoppler sidebands in the carrier frequency spectrum which are thenshifted into the same region of the baseband output from thedemodulator. The Rx signal may be expressed as follows:

$\begin{matrix}{{S(t)} = {\underset{\underset{Carrier}{︸}}{A_{0}{\cos\left( {{\omega_{0}t} + \varphi_{0}} \right)}} + \underset{\underset{Forward}{︸}}{A_{f}{\cos\left( {{\omega_{0}t} + {\omega_{f}\; t} + \varphi_{f}} \right)}} + \underset{\underset{Reverse}{︸}}{A_{r}{\cos\left( {{\omega_{0}t} - {\omega_{r}t} + \varphi_{r}} \right)}}}} & {{Eq}.\mspace{14mu} 1}\end{matrix}$where A, ω, and φ represent the amplitude, angular frequency and phaseof each signal component, and the subscripts 0, f, and r signify thecarrier, forward (toward the transducer probe), and reverse (away fromthe probe) components. The system of FIG. 1 is unable to distinguishbetween the forward and reverse sidebands.

To determine the direction in which the object is moving (i.e., towardsor away from the transceiver), a more sophisticated demodulationtechnique is required. Some examples of directional demodulationtechniques include single side-band demodulation (SSB), heterodynedemodulation, and quadrature demodulation.

FIG. 2 illustrates a basic Doppler measurement system that utilizesquadrature demodulation to obtain directional information on themovement of an object. The system of FIG. 2 is similar to that of FIG.1, but the demodulator includes two mixers 22 and 26 which mix the Rxsignal with an in-phase (“I”) clock signal from the master oscillator,and a quadrature (“Q”) clock signal that is phase shifted (90 degrees)from the I clock signal. Thus, a quadrature demodulator is also referredto as an I/Q demodulator.

After high-pass and low-pass filtering in filter stages 24 and 28, thedemodulated signals I′(t) and Q′(t) contain only the Doppler components:

$\begin{matrix}{{I^{\prime}(t)} = {{\frac{1}{2}A_{f}{\cos\left( {{\omega_{f}t} + \phi_{f}} \right)}} + {\frac{1}{2}A_{r}{\cos\left( {{\omega_{r}t} + \phi_{r}} \right)}}}} & {{Eq}.\mspace{14mu} 2} \\{{Q^{\prime}(t)} = {{{- \frac{1}{2}}A_{f}{\sin\left( {{\omega_{f}t} + \phi_{f}} \right)}} + {\frac{1}{2}A_{r}{\sin\left( {{\omega_{r}t} + \phi_{r}} \right)}}}} & {{Eq}.\mspace{14mu} 3}\end{matrix}$Further processing extracts the forward and reverse components from theI′(t) and Q′(t) signals.

Transducers having multiple transmit and/or receive elements may be usedto improve the basic performance of a Doppler measurement system, or toprovide additional functionality. For example, multiple elementtransmitters and receivers may be used to implement beamformingtechniques in which the maximum transmit and receive strength of thetransducer are pointed in the direction of the object to be measured.The elements are arranged in an array that is scanned by a beamformingnetwork so that the signal to or from each element is phase (or time)shifted relative to the other elements. In the transmit path, abeamforming network shifts the transmit signal by different phaseamounts as it distributes the signal to the different elements in thescanning process. In the receive path, a beamforming network shifts thereceive signals by different amounts as the array is scanned so thesignals from the individual elements can be phase (or time) aligned andthen combined (summed) to form a single receive signal. The summedsignal level theoretically thereby increases by N where N is the numberof array elements. If noise is uncorrelated, it increases by √{squareroot over (N)}, thereby increasing the summed signal-to-noise ratio(SNR) by N/√{square root over (N)}=√{square root over (N)}.

FIG. 3 illustrates the basic principle of using an array of elements forbeamforming. In this example, which is generic to both transmit andreceive scanning, the signals for array elements A₁ through A_(N) areshown shifted by various amounts relative to time t=0. The amount eachsignal is shifted during the scanning process determines the angle Θ ofthe beam. Numerous scanning techniques have been developed includingtime-delay scanning and phase scanning; if done in the analog domain, itis called analog beamforming, if done in the digital domain it is calleddigital beamforming (DBF). There are two general ways to beamform. Oneis to align and sum the signals at the RF frequency as shown in FIG. 4.The other is at baseband, in which case each channel requiresdownconversion and filtering before summation.

FIG. 4 illustrates a beamforming system for time-delay scanning an arrayof receive elements in an ultrasound system that utilizes RF summationfollowed by quadrature demodulation. The receiver array 30 includesmultiple receive crystals A₁ through A_(N) (where N is typically a powerof 2) which convert the received ultrasound waves into electricalsignals that are amplified by low noise amplifiers (LNAs) 34. Theoutputs from the LNAs are converted to current-mode signals by V-Iconverters 36. A multiplexer array 38 selectively connects thecurrent-mode signals to the input taps of a delay line circuit 40. Thedelay line circuit imparts a different time delay to each input signaldepending on which input tap it is applied to. All of the delayed inputsignals are then summed at node N1 to produce a composite receive signalwhich is then quadrature demodulated by multipliers 42 and 44 inresponse to the I clock signal cos(ω₀t) and the Q clock signal sin(ω₀t).The resulting signals are then band-pass filtered and converted todigital form for further processing.

Although the system of FIG. 4 may provide effective beamforming anddemodulation, it generally requires a very high performance demodulatorwhich drives up the system power and cost. Also, since delay linecircuits are generally cumbersome to add to each individual receiveelement, the system includes multiplexer array 38 to allow the use of asingle delay line circuit where the currents are routed to theappropriate tap to insure coherent addition at node N1. In a DBF system,a separate memory for each channel functions as the “delay line”.Selecting the appropriate memory locations allows multiple beams to beformed at the same time. However, continuous wave (CW) Doppler signalstend to have dynamic ranges that are too large to be processed by a DBFsystem. In addition, multiplexer arrays tend to require extensive spaceon an integrated circuit.

FIG. 5 illustrates a beamforming system for phase scanning an array ofreceive elements in an ultrasound system that utilizes quadraturedemodulation. The system of FIG. 5 includes an array of receive crystalsA₁ through A_(N) and low noise amplifiers (LNAs) 34. The output fromeach LNA is demodulated by an I mixer 46 and a Q mixer 48 in response toI and Q clock signals (which are not shown here for simplicity). Phaserotators 50 and 52 are placed in series after the mixers to phase shiftthe I_(1 . . . N) and Q_(1 . . . N) signals in response to the phaseselect signals Φ₁ through Φ_(N). The I_(1 . . . N) and Q_(1 . . . N)outputs are summed to generate I(t) and Q(t) which may then be convertedto voltage mode signals, band-pass filtered, etc.

Although the system of FIG. 5 requires a complete demodulator for eachreceiver, each of the demodulators is generally subject to lessdemanding criteria than the demodulator required for the system in FIG.4 since they are located before beamformation. Another salient featureis that the phase rotation is performed at baseband which is generallyless demanding than operations performed at the full RF frequencies ofthe input signals. The system in FIG. 5 is expected to require lesscircuit board space than the system of FIG. 4.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 illustrates a prior art system for measuring Doppler frequencyshift.

FIG. 2 illustrates a prior art Doppler measurement system that utilizesquadrature demodulation to obtain directional information on themovement of an object.

FIG. 3 illustrates the basic principle of using an array of elements forbeamforming.

FIG. 4 illustrates a prior art beamforming system with time-delayscanning and quadrature demodulation.

FIG. 5 illustrates a prior art beamforming system with phase scanningand quadrature demodulation.

FIG. 6 illustrates an embodiment of a technique for providing phaseshift in a single channel I/Q demodulator according to the inventiveprinciples of this patent disclosure.

FIG. 7 is a phasor diagram of discrete phase rotation values.

FIG. 8 illustrates an embodiment of a system for providing phase shiftin discrete steps in a quadrature demodulation system according to theinventive principles of this patent disclosure.

FIG. 9 illustrates an additional embodiment of a system for providingphase shift in discrete steps according to the inventive principles ofthis patent disclosure.

FIG. 10 illustrates another embodiment of a system for providing phaseshift in discrete steps according to the inventive principles of thispatent disclosure.

FIG. 11 illustrates an embodiment of one possible circuit implementationof the system of FIG. 10 according to the inventive principles of thispatent disclosure.

FIG. 12 illustrates a fully differential embodiment of a circuitimplementation of a system for providing phase shift according to theinventive principles of this patent disclosure.

FIG. 13 illustrates an alternative embodiment of a system for providingphase shift in discrete steps according to the inventive principles ofthis patent disclosure.

FIG. 14 illustrates an embodiment of a system for providing phase shiftin a quadrature demodulator according to the inventive principles ofthis patent disclosure.

FIG. 14A is an example of a suitable H-bridge circuit.

FIG. 15 illustrates an embodiment of a system for providing discretephase shift in a quadrature demodulator according to the inventiveprinciples of this patent disclosure.

FIG. 16 illustrates another embodiment showing some exampleimplementation details of a system according to the inventive principlesof this patent disclosure.

FIG. 17 illustrates another embodiment of a demodulation system havingphase rotation according to the inventive principles of this patentdisclosure.

FIG. 18 illustrates one possible approach to implementing thedemodulation system of FIG. 17 according to the inventive principles ofthis patent disclosure.

FIG. 19 illustrates another approach to implementing the demodulationsystem of FIG. 17 according to the inventive principles of this patentdisclosure.

FIG. 20 illustrates an embodiment of a quadrant select circuit accordingto the inventive principles of this patent disclosure.

DETAILED DESCRIPTION

This patent disclosure includes numerous inventions relating to phaseshift in demodulator systems. These inventive principles haveindependent utility and are independently patentable. In some cases,additional benefits are realized when some of the principles areutilized in various combinations with one another, thus giving rise toyet more patentable inventions. These principles may be realized incountless different embodiments. Although some specific details areshown for purposes of illustrating the preferred embodiments, othereffective arrangements can be devised in accordance with the inventiveprinciples of this patent disclosure.

FIG. 6 illustrates an embodiment of a technique for providing phaseshift in a single channel I/Q demodulator. The embodiment of FIG. 6 isan analog embodiment derived from a digital mechanization of a Fouriertransform disclosed in Radar System Design & Analysis, S. A.Hovanessian, pp. 264-268, Artech House, 1984. The input signal isapplied to I and Q mixers 46 and 48 which generate baseband signals I′and Q′ by mixing the input signal with I and Q clock signals,respectively. Since the input signal is typically at a frequency that ismuch higher than the modulation signal, it will be referred to here asthe RF signal. The I and Q clock signals are typically generated by alocal oscillator (or master oscillator) operating at the same RFfrequency and will be referred to as LOI and LOQ. The inventiveprinciples, however, are not limited to systems having any particularclock sources, or that operate at any particular frequency.

Phase rotation of the I′ and Q′ signals is accomplished by mixers 54,56, 58 and 60, and a pair of summing circuits 62 and 64. The mixersgenerate a group of four component signals by multiplying each of the I′and Q′ signals by first and second phase select signals sin(φ) andcos(φ) which are scalars that weight signals I(t) and Q(t). The fourcomponent signals are then recombined by summing circuits 62 and 64 togenerate the I(t) and Q(t) output signals which may then be filtered andfurther processed in any suitable manner.

Although the system of FIG. 6 performs the phase rotation (shifting)operation at baseband, it may require a complex arrangement of analogmixers. Note that a complete system may be required for each channel ina beamforming system.

In many applications, continuous phase rotation is not required. Thatis, the phase rotation may be implemented in discrete steps. Forexample, in continuous wave (CW) Doppler ultrasound for medical imaging,adequate beamforming capability may be obtained by dividing the 360° (2πradians) of possible phase rotation into eight or sixteen discrete phaseshifts; 45° and 22.5° each, respectively. In this case, the complexmixer arrangement may be simplified into a simpler arrangement in whichI′ and Q′ are weighted by predetermined values since the phase selectsignals sin(φ) and cos(φ) only take on the discrete values shown in thephasor diagram in FIG. 7.

FIG. 8 illustrates an embodiment of a system for providing phase shiftin discrete steps in a quadrature demodulation system according to theinventive principles of this patent disclosure. The embodiment of FIG. 8includes I and Q mixers 46 and 48 which mix the RF signal with the I andQ clock signals LOI and LOQ to generate the I′ and Q′ signals. Ratherthan an array of additional mixers, however, the embodiment of FIG. 8includes phase rotators 66 and 68 which generate a series of componentsignals by multiplying each of the I′ and Q′ signals by pairs ofweighting scalars which in this case are sine and cosine values for theappropriate phase angle φ shown in FIG. 8. A phase select signal, showngenerically as Φ, (and which may be a collection of one or more digitaldecoding signals) determines which weighting functions are applied. Notethat “C” in FIG. 8 incorporates the minus sign before the summer in FIG.6, therefore the equation for Q(t)=D+C in FIG. 8 compared to D-Ĉ in FIG.6.

The weighted component signals may then be recombined by summingcircuits 62 and 64 to generate the I(t) and Q(t) output signals whichmay then be filtered and further processed in any suitable manner, or tofurther simplify the embodiment of FIG. 8, a quadrant select circuit 70may be included to reduce the number of weighting values in each of thephase rotators. To understand how quadrant selection may simplify thesystem, reference is made to the phasor diagram of FIG. 7. All of themagnitudes of weighting values are present in the first quadrant(Quadrant 1) of FIG. 7. The remaining values may be obtained byinverting (multiplying by −1) and/or swapping I and Q signals. Thequadrant select circuit 70 performs the inversion and/or swappingfunctions in response to all or a portion of the phase select signal Φ.

In a practical realization, the LOI and LOQ signals are generated by adivider that is driven by a reference clock that operates, e.g., at fourtimes the frequency of the LOI and LOQ signals. Also in a practicalrealization, the quadrant select circuit includes a reset input(preferably made available as an external chip input) that may be usedto ensure correct phase rotation at power-up or during a reset event.For example, if the system of FIG. 8 is used to implement multipledemodulator channels in a beamforming system, the LO divider may powerup or reset shifted 180 degree in one channel versus 0 degrees inanother. Thus, a 45 degree phase shift may end up as a 45+180=225 degreephase shift. A reset input may be used to eliminate such problems.

FIGS. 9 and 10 illustrate additional embodiments of systems forproviding phase shift in discrete steps according to the inventiveprinciples of this patent disclosure. The embodiments of FIGS. 9 and 10illustrate two different approaches to implementing the phase rotatorsof FIG. 8. In the embodiment of FIG. 9, a first set of weighting circuitpairs 72A-D generate four pairs of sine/cosine weighted component signalpairs by multiplying the I′ signal by the sine and cosine values of22.5, 45, 67.5 and 90 degrees. Select circuit 73 passes a selected pairof signals through to summing circuits 62 and 64.

On the Q side of the system, a second set of weighting circuit pairs74A-D and a second select circuit 76 generate a series of sine/cosineweighed component signal pairs from the Q′ signal. The outputs from thesecond select circuit are applied to the summing circuits 62 and 64which, together with the weighted signals from the I side of the systemgenerate the I(t) and Q(t) signals. For any desired phase rotation, theselect circuits select the signal pairs from corresponding pairs ofweighting circuits on the I and Q sides of the system. That is, theoutputs from weighting circuit pair 72A are selected at the same time asthe outputs from weighting circuit pair 74A. A quadrant select circuitmay then be used to invert and/or swap I and Q signals to obtain thefinal output signals to obtain full four-quadrant operation.

The embodiment of FIG. 10 essentially reverses the order of theweighting and select circuits. Here, the first select circuit 76selectively passes the I′ signal to one of the first set of weightingcircuit pairs 78A-D. Likewise, the second select circuit 80 selectivelypasses the Q′ signal to one of the second set of weighting circuit pairs82A-D. This may effectively enable only the relevant pair of weightingcircuits in each phase rotator. The outputs of the weighting circuitsare connected in parallel to the summing circuits 62 and 64. An inverter84 provides the proper negative sign for the I′ sin(φ) signal needed inthe generation of the Q(t) output signal. Again, a quadrant selectcircuit may then be used to invert and/or swap I and Q signals to obtainthe final output signals to obtain full four-quadrant operation.

Current Mirror Weighting Circuits

FIG. 11 illustrates an embodiment of a circuit implementation of thesystem of FIG. 10 according to the inventive principles of this patentdisclosure. A transconductance (gm) cell 86 converts the input signal RFto a current mode signal which is applied to the input of a dual-outputcurrent mirror 90. One output drives the I side of the system, while theother drives the Q side. The current mirror is shown having a currentratio of 1:M:M meaning each of the output currents is M times the inputcurrent. Any suitable value of current ratio may be used, just as anysuitable bias (or “tail”) current I_(T) may be used to bias the gm cell.Mixer core 46 mixes the current mode reflection of the input signal withthe I (in-phase) local oscillator signal LOI to generate the I′ signal.

Select circuit 76 steers the I′ signal to one of four dual-outputcurrent mirrors 78A-D each of which has a current ratio of 1:A:Ĉ. On theother side of the system, select circuit 80 steers the Q′ signal to oneof four dual-output current mirrors 82A-D each of which has a currentratio of 1:B:D. The values of A, B, Ĉ and D for the current mirrors areselected to provide the proper weighting based on the selected amount ofphase rotation. Thus, the embodiment of FIG. 11 uses ratioed currentmirrors to implement the weighting functions.

The current mirror outputs, which are the weighted component signals,are summed by summing circuits 62 and 64. An inverter 84 is used on theC component signals to provide the correct polarity for C going into thesumming circuit that generates the Q(t) signal. The outputs from thesumming circuits are the I(t) and Q(t) signals which may then bequadrant selected, filtered, and/or further processed as may be needed.

For simplicity, the embodiment of FIG. 11 is shown as a single-endedimplementation. FIG. 12 illustrates some possible implementation detailsfor a fully differential embodiment. Only the I side of the system isshown. The Q side would have essentially the same structure except forthe inversion indicated at 84. The select circuit 76 is realized as aseries of cascode transistors that route the I′ signal to theappropriate current mirrors in response to the phase select signalsSEL1-4. Current mirrors 92 and 94 convert the final A and C componentsignals to single-ended form. Because the output signals are currentmode, the summing circuits may be implemented tying the outputstogether. The differential structure allows the inverter 84 to beimplemented by simply switching inputs to the C current mirror 94. Toimplement the “times 0” weight, one output of current mirror 78A issimply omitted.

FIG. 13 illustrates an alternative embodiment of a system for providingphase shift in discrete steps according to the inventive principles ofthis patent disclosure. The embodiment of FIG. 13 eliminates the currentmirror (1:M:M) and thereby reduces power and noise. This simplifiedinput signal path may result in reduced noise at the input, albeit, atthe possible expense of power supply headroom. Only the I side of thesystem is shown. The Q side would have essentially the same structure.

Demodulator with Stacked Mixers

FIG. 14 illustrates an embodiment of a system for providing phase shiftin a quadrature demodulator according to the inventive principles ofthis patent disclosure. The embodiment of FIG. 14 employs a “stacked”structure of mixers. That is, the mixer cores 46 and 48 are arranged inseries between the power supplies. An H-bridge 95 (which implements a gmstage) is arranged between the mixer cores and is utilized as the inputstage to provide tightly integrated, fully differentialvoltage-to-current conversion of the input signal. An example of asuitable H-bridge circuit is illustrated in FIG. 14A. The select circuit80, mixer core 48, and weighting circuits 82A-D on the Q side of thesystem may be implemented with opposite polarity to those on the I sideof the system so that the entire system of input stage, mixer cores,select circuits and weighting circuits may be arranged in a stackedconfiguration. Although the phase rotators are shown here as discrete,selected weighting circuits, the stacked mixer core principles are notlimited to use with any particular type of input stage or technique forphase rotation.

Integrated Mixers and Select Circuits

FIG. 15 illustrates an embodiment of a system for providing discretephase shift in a quadrature demodulator according to the inventiveprinciples of this patent disclosure. In the embodiment of FIG. 15, themixing and phase selection functions are combined in a single circuit.On the I side of the system, a first mixer/selector circuit 96 includesa set of mixers 98A-D that can be selectively enabled in response to aphase select signal Φ. Any selected mixer combines the RF input signalwith the LOI signal and directs the resulting signal to one of theweighting circuit pairs 78A-D. On the Q side, a second mixer/selectorcircuit 100 includes a second set of mixers 102A-D that perform asimilar mixing and selection operation. The component signal outputsfrom the weighting circuits may then be recombined, quadrant selected,and/or further processed in any suitable manner such as that describedwith reference to FIG. 10.

FIG. 16 illustrates another embodiment showing some exampleimplementation details of a system according to the inventive principlesof this patent disclosure. The embodiment of FIG. 16 combines theprinciples of integrated mixer/selector circuits with the principles ofstacked mixer cores described above. An H-bridge 95 converts adifferential voltage input signal to differential current mode signalsthat drive the paralleled inputs of mixer cores 98A-D and 102A-D thatare stacked with the H-bridge. The mixer cores in the I side of thesystem are enabled in response to the phase select signal Φ byselectively applying the LOI signal to the mixer cores. This, in turn,selects which of the weighting circuit pairs 78A-D (shown here asdual-output current mirrors) receives the output signal from a selectedmixer. This arrangement may provide a tightly arranged input thatreduces noise, while eliminating the power supply headroom penalty thatmay result from having the select circuits (cascodes) separate from themixers.

Pre-Weighted Input to Mixers

FIG. 17 illustrates another embodiment of a demodulation system havingphase rotation according to the inventive principles of this patentdisclosure. The embodiment of FIG. 17 places phase rotators 104 and 106upstream from the mixers so that the mixer inputs are preweighted. The Iside phase rotator 104 generates a pair of weighted signals bymultiplying the RF input signal by selected pairs of weighting scalars,which in this case are sine and cosine functions. A phase select signal,shown generically as Φ, (and which may be a collection of one or moredigital decoding signals) determines which weighting is applied. Each ofthe weighted signals is then mixed with the in-phase local oscillatorsignal LOI by a corresponding one of mixers 108 and 110 to generate Aand C component signals.

The Q side phase rotator 106 has a similar structure and generatesanother selected pair of weighted signals which are then mixed with thequadrature local oscillator signal LOQ by corresponding mixers 112 and114 to generate B and D component signals. A pair of summing circuits116 and 118 recombine the component signals to generate the I(t) andQ(t) output signals which may then be quadrant selected, filtered and/orfurther processed in any suitable manner.

FIG. 18 illustrates one possible approach to implementing thedemodulation system of FIG. 17 according to the inventive principles ofthis patent disclosure. In the embodiment of FIG. 18, the I side phaserotator 104 is realized as a set of weighting circuit pairs 120A-D and aselect circuit 122 which selects the outputs from one of the weightingcircuit pairs in response to the phase select signal Φ. The Q side phaserotator 106 has a similar structure and provides weighted RF inputs tomixers 112 and 114 which receive the LOQ signal.

Although the embodiments of FIGS. 8 and 9 may appear to provide the sameresults as the embodiments of FIGS. 17 and 18 in the abstractmathematical sense, the arrangement of components may have importantramifications for the real world performance of an actual circuitimplementation. For example, the embodiment illustrated in FIG. 11 maybe seen as a “direct” implementation of the system of FIG. 8. It hascertain operating characteristics such as noise levels, powerconsumption, power supply headroom, etc., that may be inherentlyconstrained by the arrangement of components. These constraints may bequite different from those relating to implementations of theembodiments of FIGS. 17 and 18, an example of which is described withreference to FIG. 19.

FIG. 19 illustrates another approach to implementing the demodulationsystem of FIG. 17 according to the inventive principles of this patentdisclosure. In the embodiment of FIG. 19, the weighting circuits 120A-Dand 124A-D are implemented as transconductance (gm) cells having biascurrents and degeneration resistors that are sized to provide therequisite amount of gain so as to weight the RF input signalappropriately. The outputs from the cosine weighted gm cells on the Iside are connected in parallel to one input of mixer core 108 whichgenerates the A component signal by mixing the weighted RF input withthe LOI signal. The outputs from the sine weighted gm cells areconnected in parallel to one input of mixer core 110 which generates theC component signal. A weight of “zero” is achieved by simply omittingthe gm cell. Note that the C component signal is inverted (−1) byswapping the IF outputs (Ĉ) of the mixer cores at 84. A gm cell isselected by enabling its corresponding bias current source in responseto one of the phase select signals SEL1 through SEL4. Thus, the selectfunction is integrated into the weighting circuits. Alternatively, aseries of cascode transistors may be interposed between the gm cells andthe mixers. The Q side of the system has a similar arrangement thatgenerates the B and D component signals in response to the LOQ signal.Because the component signals are in current mode, the summing circuits116 and 118 may be implemented as simple summing nodes (FIG. 20).

The A, B, C and D component signals, which in this example aredifferential current mode signals, may be applied to a quadrant selectcircuit such as that illustrated in FIG. 20. Two banks of switchedcascode transistors 126 and 128 select the appropriate pairs ofcomponent signals in response to quadrant select signals QS1-4. Theselected signal pairs are passed through to I and Q side current mirrors130 and 132, respectively, which convert the differential currents intosingle-ended I(t) and Q(t) signals.

The embodiments of FIGS. 19 and 20 may provide a beneficial tradeoff ofdynamic range and power consumption, especially when used in beamformingsystems for CW Doppler ultrasound and radar. Since only a few of theinput gm cells may be selected at any given time, power consumption maybe reduced. Also, the arrangement of gm cells provides a tightly coupledinput stage that may extend the dynamic range by lowering the noisefloor. The overlapping operation of the gm cells and mixers on the I andQ sides may reduce the amount of power supply headroom required by thesystem.

The principles disclosed above can be realized in countless differentembodiments. Only the preferred embodiments have been described.Although some specific details are shown for purposes of illustratingthe preferred embodiments, other equally effective arrangements can bedevised in accordance with the inventive principles of this patentdisclosure. For example, some transistors have been illustrated asbipolar junction transistors (BJTs), but CMOS and other types of devicesmay be used as well. Likewise, some signals and mathematical values havebeen illustrated as voltages or currents, but the inventive principlesof this patent disclosure are not limited to these particular signalmodes. In the examples above embodiments having 16 discrete phase stepsare illustrated, but the inventive principles may be applied to systemshaving a different number of steps. Since the embodiments describedabove can be modified in arrangement and detail without departing fromthe inventive concepts, such changes and modifications are considered tofall within the scope of the following claims.

The invention claimed is:
 1. An analog demodulation method comprising:mixing an input signal with an in-phase signal; mixing the input signalwith a quadrature signal; and preweighting the input signal in responseto a phase select signal before mixing with the in-phase and quadraturesignals to provide selectable phase shift.
 2. A method according toclaim 1 further comprising summing the results from mixing.
 3. A methodaccording to claim 2 where preweighting comprises preweighting bydiscrete amounts.
 4. A method according to claim 3 further comprisingquadrant selecting the results from summing.
 5. A method according toclaim 1 where preweighting comprises: weighting the input signal by afirst scalar; and simultaneously weighting the input signal by a secondscalar.
 6. A method according to claim 5 where: the first scalarcomprises a cosine value to produce a first cosine weighted output; andthe second scalar comprises a sine value to produce a first sineweighted output.
 7. A method according to claim 6 where mixing with thein-phase signal comprises: mixing the first cosine weighted output withthe in-phase signal to produce a first in- phase component; and mixingthe first sine weighted output with the in-phase signal to produce asecond in- phase component.
 8. A method according to claim 5 wherepreweighting further comprises: weighting the input signal by a thirdscalar; and weighting the input signal by a fourth scalar.
 9. A methodaccording to claim 8 where: the first scalar comprises a cosine value toproduce a first cosine weighted output; the second scalar comprises asine value to produce a first sine weighted output; the third scalarcomprises a sine value to produce a second sine weighted output; thefourth scalar comprises a cosine value to produce a second cosineweighted output.
 10. A method according to claim 9 where: mixing withthe in-phase signal comprises: mixing the first cosine weighted outputwith the in-phase signal to produce a first in-phase component, andmixing the first sine weighted output with the in-phase signal toproduce a second in-phase component; and mixing with the quadraturesignal comprises: mixing the second sine weighted output with thequadrature signal to produce a first quadrature component, and mixingthe second cosine weighted output with the quadrature signal to producea second quadrature component.
 11. A method according to claim 10further comprising: summing the first in-phase component with the firstquadrature component to produce an in-phase output; and summing thesecond in-phase component with the second quadrature component toproduce a quadrature output.
 12. A method according to claim 11 wherethe sine and cosine values are discrete values.
 13. A method accordingto claim 12 further comprising quadrant selecting the in-phase outputand the quadrature output.
 14. A quadrature demodulator comprising: afirst phase rotator to generate first and second weighted signals byweighting an input signal by a first scalar and a second scalar inresponse to a phase select signal; a second phase rotator to generatethird and fourth weighted signals by weighting the input signal by athird scalar and a fourth scalar in response to the phase select signal;first and second mixers to generate first and second component signalsby mixing the first and second weighted signals with an in-phase signal;and third and fourth mixers to generate third and fourth componentsignals by mixing the third and fourth weighted signals with aquadrature signal.
 15. A quadrature demodulator according to claim 14further comprising: a first summing circuit to generate an in-phaseoutput signal by summing the first and third component signals; and asecond summing circuit to generate a quadrature output signal by summingthe second and fourth component signals.
 16. A quadrature demodulatoraccording to claim 15 further comprising a quadrant select circuitcoupled to the first and second summing circuits.
 17. A quadraturedemodulator according to claim 14 where: the first phase rotatorcomprises a first set of weighting circuits to receive the input signaland a first select circuit to select the outputs from the firstweighting circuits; and the second phase rotator comprises a second setof weighting circuits to receive the input signal and a second selectcircuit to select the outputs from the second weighting circuits.
 18. Aquadrature demodulator according to claim 14 where each phase rotatorcomprises a set of transconductance cells arranged to weight the inputsignal in response to the phase select signal.
 19. A quadraturedemodulator comprising: a first weighting circuit to generate a firstweighted signal by multiplying the input signal by a first scalar, andto generate a second weighted signal by multiplying the input signal bya second scalar; a second weighting circuit to generate a third weightedsignal by multiplying the input signal by a third scalar, and togenerate a fourth weighted signal by multiplying the input signal by afourth scalar; a first mixer to generate a first component signal bymixing the first weighted signal with an in-phase signal a second mixerto generate a second component signal by mixing the second weightedsignal with the in-phase signal; a third mixer to generate a thirdcomponent signal by mixing the third weighted signal with a quadraturesignal; a fourth mixer to generate a fourth component signal by mixingthe fourth weighted signal with the quadrature signal; a first summingcircuit to generate an in-phase output signal by summing the first andthird component signals; and a second summing circuit to generate aquadrature output signal by summing the second and fourth componentsignals; where the first, second, third and fourth weighted signals allhave the same phase.